Transmitting high rate data within a MIMO WLAN

ABSTRACT

A method for transmitting high rate data within a multiple input multiple output (MIMO) wireless local area network (WLAN) begins by determining a data transmission rate. The method continues by, when the data transmission rate is between a first data rate and a second data rate, enabling two transmission paths. The method continues by, for each of the two transmission paths, determining at least one of: level of constellation, number of data subcarriers, rate code, and cyclic prefix duration.

This patent application is claiming priority under 35 USC § 119(e) toco-pending provisional patent application entitled ISSUES ON RATES ANDMODES FOR 802.11N, a provisional Ser. No. of 60/562,206, and aprovisional filing date of Apr. 14, 2004.

BACKGROUND OF THE INVENTION

1. Technical Field of the Invention

This invention relates generally to wireless communication systems andmore particularly to interoperability within a wireless communicationsystem between next generation and legacy wireless terminals.

2. Description of Related Art

Communication systems are known to support wireless and wire linedcommunications between wireless and/or wire lined communication devices;Such communication systems range from national and/or internationalcellular telephone systems to the Internet to point-to-point in-homewireless networks. Each type of communication system is constructed, andhence operates, in accordance with one or more communication standards.For instance, wireless communication systems may operate in accordancewith one or more standards including, but not limited to, IEEE 802.11(Wireless Local Area Networks “WLANs), Bluetooth, advanced mobile phoneservices (AMPS), digital AMPS, global system for mobile communications(GSM), code division multiple access (CDMA), local multi-pointdistribution systems (LMDS), multi-channel-multi-point distributionsystems (MMDS), and/or variations thereof.

Depending on the type of wireless communication system, a wirelesscommunication device, such as a cellular telephone, two-way radio,personal digital assistant (PDA), personal computer (PC), laptopcomputer, home entertainment equipment, et cetera communicates directlyor indirectly with other wireless communication devices. For directcommunications (also known as point-to-point communications), theparticipating wireless communication devices tune their receivers andtransmitters to the same channel or channels (e.g., one of the pluralityof radio frequency (RF) carriers of the wireless communication system)and communicate over that channel(s). For indirect wirelesscommunications, each wireless communication device communicates directlywith an associated base station (e.g., for cellular services) and/or anassociated access point (e.g., for an in-home or in-building wirelessnetwork) via an assigned channel. To complete a communication connectionbetween the wireless communication devices, the associated base stationsand/or associated access points communicate with each other directly,via a system controller, via the public switch telephone network, viathe Internet, and/or via some other wide area network.

For each wireless communication device to participate in wirelesscommunications, it includes a built-in radio transceiver (i.e., receiverand transmitter) or is coupled to an associated radio transceiver (e.g.,a station for in-home and/or in-building wireless communicationnetworks, RF modem, etc.). As is known, the transmitter includes a datamodulation stage, one or more intermediate frequency stages, and a poweramplifier. The data modulation stage converts raw data into basebandsignals in accordance with a particular wireless communication standard.The one or more intermediate frequency stages mix the baseband signalswith one or more local oscillations to produce RF signals. The poweramplifier amplifies the RF signals prior to transmission via an antenna.

As is also known, the receiver is coupled to the antenna and includes alow noise amplifier, one or more intermediate frequency stages, afiltering stage, and a data recovery stage. The low noise amplifierreceives inbound RF signals via the antenna and amplifies then. The oneor more intermediate frequency stages mix the amplified RF signals withone or more local oscillations to convert the amplified RF signal intobaseband signals or intermediate frequency (IF) signals. The filteringstage filters the baseband signals or the IF signals to attenuateunwanted out of band signals to produce filtered signals. The datarecovery stage recovers raw data from the filtered signals in accordancewith the particular wireless communication standard.

As is further known, the data recovery stage performs numerousoperations to recover data from the filtered signals. Such operationsinclude, for an IEEE 802.11a or IEEE 802.11g compliant receiver, guardinterval removal, fast Fourier transform, de-mapping and slicing,de-interleaving, and decoding. The decoding utilizes a channelestimation to produce the recovered data from de-interleaved data. Inaccordance with the IEEE 802.11a and/or IEEE 802.11g standard, a frameincludes a short training sequence (STS), a long training sequence(LTS), a signal field, and a plurality of data fields. The IEEE 802.11aand/or IEEE 802.11g standard further indicates that channel estimationis to be done during the long training sequence. Once the channelestimation is determined, it is used for the remainder of the frame.

Currently, next generation WLANs are being developed that will co-existwith IEEE 802.11a, IEEE 802.11b, and/or IEEE 802.11g stations (STAs) andaccess points (APs). One contemplated next generation system includes aMulti-Input-Multi-Output (MIMO) interface (802.11n). The MIMO interfaceof the next generation system must be interoperable with the legacy STAsand base stations. Interoperability requires that the legacy devicesidentify next generation transmissions and respond accordingly. Suchinteroperation includes at least two particular operations. In a firstoperation, an AP supports both legacy and next generation STAs. In asecond operation, legacy and next generation STAs share a channel, i.e.,co-channel/“overlapping” BSS. In each of these cases the Physical LayerConvergence Procedure (PLCP) header must allow an IEEE 802.11a/b/g STAto identify next generation transmissions and to deassert clear channelassessment (CCA) indication or use a protection mechanism likeRequest-to-Send/Clear-to-Send (RTS/CTS) or CTS-to-sent procedures toavoid conflict with the transmissions. In each of these cases, the nextgeneration preamble must be backwards compatible in order to allow thelegacy devices to recognize the next generation transmissions.

Next generation devices are required to meet particular data throughputrequirements. One such requirement is that an effective data rate shouldmeet or exceed 100 Mbps. This requirement translates to a 130 Mbpsrequirement at the PHY of the devices. Of course, this higher data rateis satisfied as a tradeoff in reach because transmit power is limited.In meeting these requirements, the characteristics of the transmitterand receiver must be selected. Therefore, a need exists for a nextgeneration that meets these requirements.

BRIEF SUMMARY OF THE INVENTION

The transmitting high rate data within a MIMO WLAN of the presentinvention substantially meets these needs and others. In one embodiment,a method for transmitting high rate data within a multiple inputmultiple output (MIMO) wireless local area network (WLAN) begins bydetermining a data transmission rate. The method continues by, when thedata transmission rate is between a first data rate and a second datarate, enabling two transmission paths. The method continues by, for eachof the two transmission paths, determining at least one of: level ofconstellation, number of data subcarriers, rate code, and cyclic prefixduration.

In another embodiment, a method for supporting high data rate WLANcommunications begins by determining a bandwidth of operation. Themethod continues by determining a required data throughput rate. Themethod continues by selecting a number of antennas for use in a MultipleInput Multiple Output (MIMO) baseband signal format. The methodcontinues by selecting a constellation. The method continues byoperating a MIMO WLAN transceiver according to the bandwidth ofoperation, the number of antennas, and the constellation to meet therequired data throughput rate.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

FIG. 1 is a schematic block diagram of a wireless communication systemin accordance with the present invention;

FIG. 2 is a schematic block diagram of a wireless communication devicein accordance with the present invention;

FIG. 3 is a schematic block diagram of a wireless communication devicein accordance with the present invention;

FIG. 4 is a schematic block diagram of a receiver section of thewireless communication device of FIG. 2 in accordance with the presentinvention;

FIG. 5 is a schematic block diagram of an embodiment of a basebandprocessing module of the wireless communication device of FIG. 3 inaccordance with the present invention;

FIG. 6 is a schematic block diagram of another embodiment of thebaseband processing module of the wireless communication device of FIG.3 in accordance with the present invention;

FIG. 7 is a schematic block diagram of yet another embodiment of thebaseband processing module of the wireless communication device of FIG.3 in accordance with the present invention;

FIG. 8 is a schematic block diagram of an embodiment of the basebandprocessing module of the wireless communication device of FIG. 2 inaccordance with the present invention;

FIGS. 9-11 are diagrams of various frame formats that may be processedby the baseband processing module of FIG. 8;

FIG. 12A is a diagram of a frame format that may be processed by thebaseband processing modules of FIGS. 7 and 8;

FIG. 12B is a received signal model of the signal of the frame format ofFIG. 12A;

FIG. 13 is a diagram illustrating the transmission of a preamble on aplurality of antennas that is compatible with the baseband processingmodules of both FIGS. 7 and 8;

FIG. 14 is a diagram illustrating a transmission model of the frameformat of FIG. 12A;

FIG. 15 is a diagram illustrating the manner in which a preamble of theframe format of FIG. 12A is formed for a generalized next generationMIMO transmitter and particularly for a two antenna next generation MIMOtransmitter;

FIG. 16 is a diagram illustrating the manner in which a preamble of theframe format of FIG. 12A is formed for a three antenna next generationMIMO transmitter;

FIG. 17 is a diagram illustrating the manner in which a preamble of theframe format of FIG. 12A is formed for a four antenna next generationMIMO transmitter;

FIG. 18 is a block diagram illustrating the manner in which legacy andnext generation WLAN devices interpret information contained in a headerof a frame that is backwards compatible; and

FIG. 19 is a graph illustrating the frequency characteristics of variousnext generation MIMO signal formats according to the present invention.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 is a schematic block diagram illustrating a communication system10 that includes a plurality of base stations and/or access points 12and 16, a plurality of wireless communication devices 18-32 and anetwork hardware component 34. The wireless communication devices 18-32may be laptop host computers 18 and 26, personal digital assistant hosts20 and 30, personal computer hosts 24 and 32 and/or cellular telephonehosts 22 and 28. The details of at least some of the wirelesscommunication devices will be described in greater detail with referenceto FIG. 2.

The base stations or access points 12-16 are operably coupled to thenetwork hardware 34 via local area network connections 36, 38 and 40.The network hardware 34, which may be a router, switch, bridge, modem,system controller, et cetera provides a wide area network connection 42for the communication system 10. Each of the base stations or accesspoints 12 and 16 has an associated antenna or antenna array tocommunicate with the wireless communication devices in its regionalarea, which is generally referred to as a basic service set (BSS) 11,13. Typically, the wireless communication devices register with aparticular base station or access point 12 or 16 to receive servicesfrom the communication system 10.

Typically, base stations are used for cellular telephone systems andlike-type systems, while access points are used for in-home orin-building wireless networks. Regardless of the particular type ofcommunication system, each wireless communication device includes abuilt-in radio and/or is coupled to a radio. The radio includes a highlylinear amplifier and/or programmable multi-stage amplifier as disclosedherein to enhance performance, reduce costs, reduce size, and/or enhancebroadband applications.

Wireless communication devices 22, 23, and 24 are located in an area ofthe wireless communication system 10 where they are not affiliated withan access point. In this region, which is generally referred to as anindependent basic service set (IBSS) 15, the wireless communicationdevices communicate directly (i.e., point-to-point or point-to-multiplepoint), via an allocated channel to produce an ad-hoc network.

FIG. 2 is a schematic block diagram illustrating a wirelesscommunication device that includes the host device 18-32 and anassociated radio, or station, 60. For cellular telephone hosts, theradio 60 is a built-in component. For personal digital assistants hosts,laptop hosts, and/or personal computer hosts, the radio 60 may bebuilt-in or an externally coupled component. In this embodiment, thestation may be compliant with one of a plurality of wireless local areanetwork (WLAN) protocols including, but not limited to, IEEE 802.11n.The device of FIG. 2 is a Multi-Input-Multi-Output (MIMO) device. IEEE802.11n devices are referred to herein interchangeably as nextgeneration WLAN devices while IEEE 802.11a/b/g devices are referred toherein as legacy devices, which are Multi-Input-Single-Output (MISO)devices. However, the MISO devices, illustrated in more detail withreference to FIG. 3, must be interoperable with the MIMO device of FIG.2.

As illustrated, the host device 18-32 includes a processing module 50,memory 52, radio interface 54, input interface 58 and output interface56. The processing module 50 and memory 52 execute the correspondinginstructions that are typically done by the host device. For example,for a cellular telephone host device, the processing module 50 performsthe corresponding communication functions in accordance with aparticular cellular telephone standard.

The radio interface 54 allows data to be received from and sent to theradio 60. For data received from the radio 60 (e.g., inbound data), theradio interface 54 provides the data to the processing module 50 forfurther processing and/or routing to the output interface 56. The outputinterface 56 provides connectivity to an output display device such as adisplay, monitor, speakers, et cetera such that the received data may bedisplayed. The radio interface 54 also provides data from the processingmodule 50 to the radio 60. The processing module 50 may receive theoutbound data from an input device such as a keyboard, keypad,microphone, et cetera via the input interface 58 or generate the dataitself. For data received via the input interface 58, the processingmodule 50 may perform a corresponding host function on the data and/orroute it to the radio 60 via the radio interface 54.

Radio, or station, 60 includes a host interface 62, a basebandprocessing module 64, memory 66, a plurality of radio frequency (RF)transmitters 68-72, a transmit/receive (T/R) module 74, a plurality ofantennas 82-86, a plurality of RF receivers 76-80, and a localoscillation module 100. The baseband processing module 64, incombination with operational instructions stored in memory 66, executedigital receiver functions and digital transmitter functions,respectively. The digital receiver functions include, but are notlimited to, digital intermediate frequency to baseband conversion,demodulation, constellation demapping, decoding, de-interleaving, fastFourier transform, cyclic prefix removal, space and time decoding,and/or descrambling. The digital transmitter functions include, but arenot limited to, scrambling, encoding, interleaving, constellationmapping, modulation, inverse fast Fourier transform, cyclic prefixaddition, space and time encoding, and/or digital baseband to IFconversion. The baseband processing modules 64 may be implemented usingone or more processing devices. Such a processing device may be amicroprocessor, micro-controller, digital signal processor,microcomputer, central processing unit, field programmable gate array,programmable logic device, state machine, logic circuitry, analogcircuitry, digital circuitry, and/or any device that manipulates signals(analog and/or digital) based on operational instructions. The memory 66may be a single memory device or a plurality of memory devices. Such amemory device may be a read-only memory, random access memory, volatilememory, non-volatile memory, static memory, dynamic memory, flashmemory, and/or any device that stores digital information. Note thatwhen the processing module 64 implements one or more of its functionsvia a state machine, analog circuitry, digital circuitry, and/or logiccircuitry, the memory storing the corresponding operational instructionsis embedded with the circuitry comprising the state machine, analogcircuitry, digital circuitry, and/or logic circuitry.

In operation, the radio 60 receives outbound data 88 from the hostdevice via the host interface 62. The baseband processing module 64receives the outbound data 88 and, based on a mode selection signal 102,produces one or more outbound symbol streams 90. The mode selectionsignal 102 will indicate a particular mode as are illustrated in themode selection tables, which appear at the end of the detaileddiscussion. For example, the mode selection signal 102 may indicate afrequency band of 2.4 GHz, a channel bandwidth of 20 or 22 MHz and amaximum bit rate of 54 megabits-per-second or greater, e.g., 122 MBPS.In this general category, the mode selection signal will furtherindicate a particular rate. In addition, the mode selection signal mayindicate a particular type of modulation, which includes, but is notlimited to, Barker Code Modulation, BPSK, QPSK, CCK, 16 QAM and/or 64QAM.

The baseband processing module 64, based on the mode selection signal102 produces the one or more outbound symbol streams 90 from the outputdata 88. For example, if the mode selection signal 102 indicates that asingle transmit antenna is being utilized for the particular mode thathas been selected, the baseband processing module 64 will produce asingle outbound symbol stream 90. Alternatively, if the mode selectsignal indicates 2, 3 or 4 antennas, the baseband processing module 64will produce 2, 3 or 4 outbound symbol streams 90 corresponding to thenumber of antennas from the output data 88.

Depending on the number of outbound streams 90 produced by the basebandmodule 64, a corresponding number of the RF transmitters 68-72 will beenabled to convert the outbound symbol streams 90 into outbound RFsignals 92. The transmit/receive module 74 receives the outbound RFsignals 92 and provides each outbound RF signal to a correspondingantenna 82-86.

When the radio 60 is in the receive mode, the transmit/receive module 74receives one or more inbound RF signals via the antennas 82-86. The T/Rmodule 74 provides the inbound RF signals 94 to one or more RF receivers76-80. The RF receiver 76-80, which will be described in greater detailwith reference to FIG. 4, converts the inbound RF signals 94 into acorresponding number of inbound symbol streams 96. The number of inboundsymbol streams 96 will correspond to the particular mode in which thedata was received. The baseband processing module 60 receives theinbound symbol streams 90 and converts them into inbound data 98, whichis provided to the host device 18-32 via the host interface 62. For afurther discussion of an implementation of the radio, or station, 60refer to co-pending patent application entitled WLAN TRANSMIITER HAVINGHIGH DATA THROUGHPUT, having an attorney docket number of BP 3516, and aprovisional filing date of Feb. 19, 2004 and co-pending patentapplication entitled WLAN RECEIVER HAVING AN ITERATIVE DECODER, havingan attorney docket number of BP 3529 and a provisional filing date ofFeb. 19, 2004.

In one embodiment of the radio 60, a method for transmitting high ratedata within a multiple input multiple output (MIMO) wireless local areanetwork (WLAN) begins by determining a data transmission rate. Themethod continues by, when the data transmission rate is between a firstdata rate and a second data rate, enabling two transmission paths. Themethod continues by, for each of the two transmission paths, determiningat least one of: level of constellation, number of data subcarriers,rate code, and cyclic prefix duration.

In another embodiment of the radio 60, a method for supporting high datarate WLAN communications begins by determining a bandwidth of operation.The method continues by determining a required data throughput rate. Themethod continues by selecting a number of antennas for use in a MultipleInput Multiple Output (MIMO) baseband signal format. The methodcontinues by selecting a constellation. The method continues byoperating a MIMO WLAN transceiver according to the bandwidth ofoperation, the number of antennas, and the constellation to meet therequired data throughput rate.

As one of average skill in the art will appreciate, the wirelesscommunication device of FIG. 2 may be implemented using one or moreintegrated circuits. For example, the host device may be implemented onone integrated circuit, the baseband processing module 64 and memory 66may be implemented on a second integrated circuit, and the remainingcomponents of the radio 60, less the antennas 82-86, may be implementedon a third integrated circuit. As an alternate example, the radio 60 maybe implemented on a single integrated circuit. As yet another example,the processing module 50 of the host device and the baseband processingmodule 64 may be a common processing device implemented on a singleintegrated circuit. Further, the memory 52 and memory 66 may beimplemented on a single integrated circuit and/or on the same integratedcircuit as the common processing modules of processing module 50 and thebaseband processing module 64.

FIG. 3 is a schematic block diagram illustrating a wirelesscommunication device that includes the host device 18-32 and anassociated radio 61. For cellular telephone hosts, the radio 61 is abuilt-in component. For personal digital assistants hosts, laptop hosts,and/or personal computer hosts, the radio 61 may be built-in or anexternally coupled component. The host device 18-32 operates asdiscussed above with reference to FIG. 1. The WLAN device of FIG. 3 mayoperate in compliance with one or more of the IEEE 802.11a/b/g operatingstandards. As distinguished from the MIMO device of FIG. 2, the deviceof FIG. 3 is a MISO device.

Radio 61 includes a host interface 62, baseband processing module 64, ananalog-to-digital converter 111, a filter module 109, an IF mixing downconversion stage 107, a receiver filter 101, a low noise amplifier 103,a transmitter/receiver switch 73, a local oscillation module 74, memory66, a digital transmitter processing module 76, a digital-to-analogconverter 78, a filter module 79, an IF mixing up conversion stage 81, apower amplifier 83, a transmitter filter module 85, and an antenna 86.The antenna 86 may be a single antenna that is shared by the transmitand receive paths as regulated by the Tx/Rx switch 73, or may includeseparate antennas for the transmit path and receive path. The antennaimplementation will depend on the particular standard to which thewireless communication device is compliant. The baseband processingmodule 64 functions as described above and performs one or more of thefunctions illustrated in FIGS. 5-19.

In operation, the radio 61 receives outbound data 88 from the hostdevice via the host interface 62. The host interface 62 routes theoutbound data 88 to the baseband processing module 64, which processesthe outbound data 88 in accordance with a particular wirelesscommunication standard (e.g., IEEE 802.11a/b/g, Bluetooth, et cetera) toproduce outbound time domain baseband (BB) signals.

The digital-to-analog converter 77 converts the outbound time domainbaseband signals from the digital domain to the analog domain. Thefiltering module 79 filters the analog signals prior to providing themto the IF up-conversion module 81. The IF up conversion module 81converts the analog baseband or low IF signals into RF signals based ona transmitter local oscillation 83 provided by local oscillation module100. The power amplifier 83 amplifies the RF signals to produce outboundRF signals 92, which are filtered by the transmitter filter module 85.The antenna 86 transmits the outbound RF signals 92 to a targeted devicesuch as a base station, an access point and/or another wirelesscommunication device.

The radio 61 also receives inbound RF signals 94 via the antenna 86,which were transmitted by a base station, an access point, or anotherwireless communication device. The antenna 86 provides the inbound RFsignals 94 to the receiver filter module 101 via the Tx/Rx switch 73.The Rx filter 71 bandpass filters the inbound RF signals 94 and providesthe filtered RF signals to the low noise amplifier 103, which amplifiesthe RF signals 94 to produce amplified inbound RF signals. The low noiseamplifier 72 provides the amplified inbound RF signals to the IF downconversion module 107, which directly converts the amplified inbound RFsignals into inbound low IF signals or baseband signals based on areceiver local oscillation 81 provided by local oscillation module 100.The down conversion module 70 provides the inbound low IF signal orbaseband signal to the filtering/gain module 68. The filtering module109 filters the inbound low IF signals or the inbound baseband signalsto produce filtered inbound signals.

The analog-to-digital converter 111 converts the filtered inboundsignals into inbound time domain baseband signals. The basebandprocessing module 64 decodes, descrambles, demaps, and/or demodulatesthe inbound time domain baseband signals to recapture inbound data 98 inaccordance with the particular wireless communication standard beingimplemented by radio 61. The host interface 62 provides the recapturedinbound data 92 to the host device 18-32 via the radio interface 54.

As one of average skill in the art will appreciate, the wirelesscommunication device of FIG. 3 may be implemented using one or moreintegrated circuits. For example, the host device may be implemented onone integrated circuit, the baseband processing module 64 and memory 66may be implemented on a second integrated circuit, and the remainingcomponents of the radio 61, less the antenna 86, may be implemented on athird integrated circuit. As an alternate example, the radio 61 may beimplemented on a single integrated circuit. As yet another example, theprocessing module 50 of the host device and the baseband processingmodule 64 may be a common processing device implemented on a singleintegrated circuit. Further, the memory 52 and memory 66 may beimplemented on a single integrated circuit and/or on the same integratedcircuit as the common processing modules of processing module 50 and thebaseband processing module 64.

FIG. 4 is a schematic block diagram of each of the RF receivers 76-80.In this embodiment, each of the RF receivers 76-80 includes an RF filter101, a low noise amplifier (LNA) 103, a programmable gain amplifier(PGA) 105, a down-conversion module 107, an analog filter 109, ananalog-to-digital conversion module 111 and a digital filter anddown-sampling module 113. The RF filter 101, which may be a highfrequency band-pass filter, receives the inbound RF signals 94 andfilters them to produce filtered inbound RF signals. The low noiseamplifier 103 amplifies the filtered inbound RF signals 94 based on again setting and provides the amplified signals to the programmable gainamplifier 105. The programmable gain amplifier further amplifies theinbound RF signals 94 before providing them to the down-conversionmodule 107.

The down-conversion module 107 includes a pair of mixers, a summationmodule, and a filter to mix the inbound RF signals with a localoscillation (LO) that is provided by the local oscillation module toproduce analog baseband signals. The analog filter 109 filters theanalog baseband signals and provides them to the analog-to-digitalconversion module 111 which converts them into a digital signal. Thedigital filter and down-sampling module 113 filters the digital signalsand then adjusts the sampling rate to produce the inbound symbol stream96.

FIG. 5 is a functional schematic block diagram of an implementation ofthe baseband processing module 64 of FIG. 3. In this embodiment, thebaseband processing module 64 is implemented to include a guard intervalremoval module 130, a fast Fourier transform (FFT) module 132, ademapping/slicing module 134, a deinterleaving module 136, a decodingmodule 138, and the channel estimation module 120. In this embodiment,the channel estimation module 120 includes an encoding module 140, aninterleaving module 142, a mapping module 144, a channel estimationmodule 146, and a channel estimation update module 148. As is furthershown, a frame 155, which may in accordance with IEEE 802.11a and/orIEEE 802.11.g, includes a short training sequence, two long trainingsequences, a service field, and a plurality of data payload sections.

The baseband processing module 64 processes the sections of frame 155sequentially. As is known, the baseband processing module 64 processesthe short training sequence to recognize the presence of a frame tobegin the determination of whether the frame is valid, and to establishinitial gain settings of the radio receiver section (e.g., the LNA gain,programmable gain amplifier gain, analog-to-digital gain, et cetera).

The baseband processing module 64 then processes the long trainingsequences to further establish the validity of frame 155 and via theguard interval removal module 130 to remove the guard intervals thatseparate the long training sequences. The fast Fourier transform module132 converts the time domain signals representing the long trainingsequences into a plurality of time domain tones 150. Thedemapping/slicing module 134 demaps the plurality of frequency domaintones 150 to produce demapped tones 152. The interleaving module 136deinterleaves the demapped tones 152 to produce deinterleaved data 154.The decoding module 138 decodes the deinterleaved data 154 to produceinbound decoded data 98.

For example, if the baseband processing module 64 is configured to becompliant with IEEE 802.11a and/or 802.11g, the inbound time domainbaseband signals are orthogonal frequency division multiplexed (OFDM)modulated signals in the 5 GHz frequency band and/or the 2.4 GHzfrequency band. The FFT module 132 converts the time domain signals intoa plurality of frequency domain tones. Each of the frequency domaintones represents a sub-carrier of a channel. As is known, in the 802.11aand/or 802.11g standard, there are 48 data sub-carriers and 4 pilotsub-carriers for a total of 52 non-zero sub-carriers of a channel. Theremaining 12 sub-carriers of the channel are zero to provide at least aportion of the guard interval. Each tone represents a modulated datathat is in accordance with one of PBSK, QPSK, 16 QAM and/or 64 QAM. Thedemapping determines the particular symbol vector for the correspondingtone which is subsequently deinterleaved via the deinterleave module136. The decoding module 138, which may be a VITERBI decoder, receivesthe symbol vectors representing the modulated data and decodes themaccordingly to recapture the bits represented by the constellationmapped symbols.

The channel estimation module 120 essentially replicates the basebandtransmit function to produce the re-mapped frequency domain tones fromdecoded data produced by the decoding module 138. As shown, the encodingmodule 140, which may be a convolutional encoder using rate 1/2, encodesthe inbound decoded data bits 98 to produce re-encoded data 156. Theencoding module 140 essentially is performing the inverse of thedecoding module 138 and is performing the same encoding function thatthe transmitting wireless communication device used to encode the datathat it transmitted to this particular receiver.

The interleaving module 142 interleaves the re-encoded data 156 toproduce reinterleaved data 158. The mapping module 144 maps thereinterleaved data 158 to a plurality of remapped frequency domain tones160. These functions are the inverse, or compliment, of the functionsperformed by the demapping/slicing module 134 and the deinterleavingmodule 136.

The channel estimation module 146 utilizes the plurality of remappedfrequency domain tones 160 and the plurality of frequency domain tones150 to produce a channel estimation 162 for the particular portion ofthe frame being processed. Accordingly, a channel estimation 162 may beproduced for the long training sequences yielding an LTS channelestimation, may be performed for the service field, which generally maybe referred to as frame information section, to produce a service fieldchannel estimation, and one or more of the data payloads may have achannel estimation 160 determined therefore.

The channel estimation update module 148 receives the channel estimation162 for the particular section of frame 155 and updates a previouschannel estimation to produce an updated channel estimation 163. As oneof average skill in the art will appreciate, the LTS channel estimationmay be derived in accordance with prior art channel estimations inwireless LAN receivers that were 802.11a and/or 802.11g compliant.

With reference to frame 155, the channel estimation module 120 generatesan initial channel estimation for the frame based on the LTS channelestimation. As the service field is being received, the channelestimation module 120 generates a service field channel estimation forthe service field. The channel estimation module 120 then updates thechannel estimation 163 for the frame based on the initial channelestimation and the newly determined service field channel estimation.When the 1^(st) data payload is received, the channel estimation module120 generates a corresponding channel estimation for this data payload.The previously updated channel estimation is then updated with the1^(st) payload channel estimation. The channel estimation module 120 maydetermine a corresponding channel estimation for each data payloadreceived and update the current channel estimation 163 accordingly.Alternatively, the channel estimation module 120 may only utilize a setof the data payload sections to determine the updating of the channelestimation 163. Which data payloads to use may be predetermined (forexample, use every 4^(th) data payload) or may be based on power of thecorresponding data payload where the energy level needs to exceed athreshold to be used for an updating of the channel estimation.

As an example of the operational of the channel estimation module 146and the channel estimation update module 148, let the received FFToutput on the K^(th) tone be:Y _(k) =Z _(k) H _(k) +V _(k)Dropping the subscript k for any tone, the equation can be rewritten as:Y=ZH+V≈CN(0, σ²)where Y is the received frame information section and/or receivedpayload section, H is the corresponding channel estimation, V representsa noise component of the received frame information section and/or thereceived payload section, and Z represents the plurality of remappedfrequency dome tones of the received frame information section and/orreceived payload section, where Z can be expressed as: Z=K_(MOD)Xtherefore: $\begin{matrix}{Y = {{\left( {Z_{i} + {jZq}} \right)\left( {{Hi} + {jHq}} \right)} + \left( {{Vi} + {jVq}} \right)}} \\{\quad{= {\left( {{Z_{i}{Hi}} - {ZqHq}} \right) + {j\left( {{ZqHi} + {ZiHq}} \right)} + {\left( {{Vi} + {jVq}} \right)\quad{therefore}\text{:}}}}} \\{{Yi} = {{ZiHi} - {ZqHq} + {Vi}}} \\{{Yq} = {{ZqHi} + {ZiHq}\quad + {V_{q}\quad{therefore}\text{:}}}}\end{matrix}$

ZiYi+ZqVq=(Zi²+Zq²)Hi+ZiVi+ZqVq, as such the channel estimation may beexpressed as: $\begin{matrix}{{\hat{H}}_{D\quad N\quad i} = {{\hat{H}}_{i} = \frac{{Z_{i}Y_{i}} + {Z_{q}V\quad q}}{Z_{i}^{2} + Z_{q}^{2}}}} \\{{\hat{H}}_{i} = {H_{i} + \frac{{Z_{i}V_{i}} + {Z_{q}V\quad q}}{Z_{i}^{2} + Z_{q}^{2}}}} \\{\quad{= \frac{{Z_{i}^{2}\sigma^{2}} + {Z_{q}^{2}\sigma^{2}}}{\left( {Z_{i}^{2} + Z_{q}^{2}} \right)^{2}}}} \\{\quad{= \frac{\sigma^{2}}{\left( {Z_{i}^{2} + Z_{q}^{2}} \right)^{2}}}} \\{\quad{= \frac{\sigma^{2}}{K_{MOD}^{2}\left( {X_{i}^{2} + X_{q}^{2}} \right)}}}\end{matrix}$

As a further example, constellation points with high energy may be usedto minimize estimation noise. For instance, consider 64 QAM, where$K_{mod} = \frac{1}{42}$$\frac{\sigma^{2}}{K_{MOD}^{2}\left( {X_{i}^{2} + X_{q}^{2}} \right)} = \frac{42\sigma^{2}}{\left( {X_{i}^{2} + X_{q}^{2}} \right)}$

From this example, channel estimation updates are done only when theconstellation energy is greater than 42. Given this premise, thefollowing constellation coordinates would give such an energy level:(X_(i), X_(q)) X_(i) ²+ X_(q) ² I1, I7 50 I3, I7 58 I5, I7 74 I7, I7 98I5, I5 50

FIG. 6 is an alternate implementation of the baseband processing module64. In this embodiment the baseband processing module 64 includes theguard interval removal module 130, the FFT module 132, thedemapping/slicing module 134, the deinterleaving module 136, thedecoding module 138, and the channel estimation module 120. In thisembodiment, the channel estimation module 120 includes the interleavingmodule 142, the mapping module 144, the channel estimation module 146and the channel estimation update module 148. Modules 130-138 functionas previously described with reference to FIG. 5 to convert inbound timedomain baseband signals into inbound decoded data 98.

In this embodiment, the channel estimation module 120 receives thedeinterleaved data 154 from module 136 via the interleaving module 142.The interleaving module 142 reinterleaves the data to producereinterleaved data 158. The mapping module 144 maps the reinterleaveddata 158 to a plurality of remapped frequency domain tones 160. Thechannel estimation module 146 and channel estimation update module 148function as previously described with reference to FIG. 5 to produce theupdated channel estimate 163.

FIG. 7 is a schematic block diagram of yet another embodiment of thebaseband processing module 64. In this embodiment, the basebandprocessing module 64 is configured to include the guard interval removalmodule 130, the FFT module 132, the demapping/slicing module 134, thedeinterleaving module 136, the decoded module 138, and the channelestimation module 120. In this embodiment, the channel estimation module120 includes the mapping module 144, the channel estimation module 146and the channel estimation update module 148. Modules 130-138 operate aspreviously described with reference to FIG. 5 to convert inbound timedomain baseband signals into inbound decoded data 98.

In this embodiment, the channel estimation module 120 receives thedemapped tones 152 via the mapping module 144. The mapping module 144maps the demapped tones 152 to tones of the OFDM modulation to produce aplurality of remapped frequency domain tones 160. The channel estimationmodule 146 and channel estimation update module 148 function aspreviously described with reference to FIG. 5 to produce the updatedchannel estimation 163.

FIG. 8 illustrates the baseband processing of a receiver in accordancewith the wireless communication device of FIG. 2. The basebandprocessing includes a space/time decoder 294, a plurality of fastFourier transform (FFT)/cyclic prefix removal modules 296-300, aplurality of symbol demapping modules 302-306, a multiplexer 308, adeinterleaver 310, a channel decoder 312, a descramble module 314, andthe channel estimation module 120. The baseband processing module mayfurther include a mode managing module 175, which produces settings 315and rate selections 311 based on mode of operation inputs 313. Thespace/time decoding module 294 receives P-inputs from the receiver pathsand produces M-output paths. In an embodiment, the space/time decodingmodule 294 multiples the input symbols of each path with a decodingmatrix that has the form of $\quad\begin{bmatrix}C_{1} & C_{2} & C_{3} & \cdots & C_{{2M} - 1} \\{- C_{2}^{*}} & C_{1}^{*} & C_{4} & \cdots & C_{2M}\end{bmatrix}$Note that the rows of the decoding matrix correspond to the number ofinput paths and the columns correspond to the number of output paths.Note that the number of M output paths of the space and time decodingmay equal the number of P-input paths of the space and time decoding orthe number of input paths P may equal M+1 paths.

The FFT/cyclic prefix removal modules 296-300 converts the M streams ofsymbols from time domain symbols to frequency domain symbols to produceM streams of frequency domain symbols. In one embodiment, the prefixremoval function removals inter-symbol interference based on a prefix.Note that, in general, a 64-point FFT will be used for 20 MHz channelsand 128-point FFT will be used for 40 MHz channels.

The symbol demapping modules 302-306 convert the frequency domainsymbols into bit streams of data. In an embodiment, each symboldemapping module maps quadrature amplitude modulated QAM symbols (e.g.,BPSK, QPSK, 16 QAM, 64 QAM, 256 QAM, et cetera) into a bit stream ofdata. Note that for IEEE 802.11(a) backward compatibility, double graycoding may be used. The multiplexer 308 combines the demapped symbolstreams into a single path. The deinterleaver 310 deinterleaves thesingle path.

The iterative decoder 312, which is described in greater detail inco-pending patent application entitled WLAN RECEIVER HAVING AN ITERATIVEDECODER having an attorney docket number of BP 3529 and a provisionalfiling date of Feb. 20, 2004, decodes the deinterleaved data to producedecoded data. The descrambler 314 descrambles the decoded data toproduce the inbound data 98. In one embodiment, the descrambler 314removes (in GF2) a pseudo random sequence from the decoded data. Apseudo random sequence may be generated from a feedback shift registerwith the generator polynomial of S(x)=x⁷+x⁴+1 to produce scrambled data.

The channel estimation module 120 may be coupled to the output of thedeinterleaving module 310 to receive deinterleaved data or it may becoupled to the output of the channel decoder 312 to receive decodeddata. If the channel estimation module 120 is coupled to receive thedecoded data it functions as previously described with reference to FIG.5. If the channel estimation module 120 receives the deinterleaved data,it functions as previously described with reference to FIG. 6.

FIG. 9 illustrates a frame 200 that may be constructed in accordancewith IEEE 802.11n when only 802.11n compliant devices are within aproximal area for a wireless communication. As shown, frame 200 includesa short training sequence (STS) 157, a plurality of supplemental longtraining sequences (suppl LTS) 201-203, and a plurality of data payloadsections 205-207. For this type of frame, the channel estimation module120 of FIG. 8 will initially generate the channel estimation based onthe LTS channel estimation as previously described with reference toFIG. 5. The channel estimation module 120 will then update the channelestimation for each channel estimation it generates for a data payloadsection. As shown, the 1^(st) data payload has a corresponding channelestimate that is used to update the LTS channel estimate to produce theupdated channel estimate. The next data payload has a correspondingchannel estimate produced for it and the corresponding channel estimateis used to update the previously updated channel estimate.

FIG. 10 illustrates a frame 202 that may be in accordance with IEEE802.11n where the communication area includes 802.11n, 802.11a and/or802.11g devices. In this instance, the frame 202 includes the shorttraining sequence (STS) 157, long training sequences (LTS) 159 & 161 inaccordance with the 802.11a and/or 802.11g standard, a service field(SIG) 163 in accordance with the 802.11a and/or 802.11g standard,supplemental long training sequences (suppl LTS) 201-203, a high dataservice field 211, and a plurality of data payload sections 205-209.Frame 202, as illustrated, includes two frame information fields: theservice field 163 and the high data service field 211.

The channel estimation module 120 of FIG. 8 generates the channelestimation by first determining the LTS channel estimation and thenupdating it with a channel estimation corresponding to the servicefield. The channel estimation module then determines a channel estimatefor the supplemental long training sequences and uses that to update thepreviously updated channel estimate. The updating of the channelestimate continues for the high data service field 211 and one or moreof the data payload fields 205-209.

FIG. 11 is another illustration of a frame 204 that may be compliantwith IEEE 802.11n for communications that include 802.11n devices,802.11a devices, 802.11b devices and/or 802.11g devices. In thisexample, the frame 204 includes a short training sequence (STS) 157, thelegacy long training sequences 1 and 2 (LTS) 159 & 161, the legacyservice field (SIG) 163, a MAC partitioning field 213, supplemental longtraining sequences (suppl LTS) 201-203, the high data service field 211and a plurality of data payload fields 205-209. The channel estimationmodule 120 of FIG. 8 determines the initial channel estimate byutilizing the LTS channel estimate. The channel estimation module 120then determines a channel estimate for each field and/or section offrame 204 and uses that channel estimate to update the previouslyupdated channel estimate. In this illustration, frame 204 includes thelegacy service field 163, the MAC partitioning field 213 and the highdata service field 211 as frame information sections.

FIG. 12A is a diagram of a portion of a frame 221 that may be processedby the baseband processing modules of FIGS. 7 and 8. The frame 221includes a short training sequence (STS) 157, a guard interval (GI) 223,two channel soundings (CS) 245 and 247, and a signal field (SIG) 163. Inone embodiment, the portion of the frame 221 is 20 microseconds (μS) induration where the STS consumes 4 μS, the GI consumes 1.6 μS, each ofthe CSs consumes 3.2 μS, and the signal field consumes 4 μS. Within theSTS, each of the symbols consumes 0.8 μS.

The STS 157 includes ten short training symbols (s₁-s₁₀) 225-243. Thechannel soundings 245 and 247 (e.g., long training in IEEE 802.11a) ofthe frame 221 satisfy 2 criteria:

-   -   1. Legacy (802.11a/g) stations can use it to decode the SIGNAL        field and get the frame length to set the clear channel        assessment (CCA) indication.    -   2. Next generation 802.11n stations can use it for (part of) the        MIMO channel estimate.

With these criteria satisfied, the channel estimation error is minimizedfor a given amount of overhead and the sequence is energy-efficient. Forunchanged SIGNAL field decoding at legacy stations, linear weighting ofthe existing long training and SIGNAL symbols at the transmitter antennainputs is used, where the same weighting is applied to the first twolong training symbols and the legacy SIGNAL field for decoding by legacystations.

FIG. 12B is a diagram of a received signal model of the signal of theframe format of FIG. 12A. As shown the received signal (X_(k)) 255 iscomprises of the transmitted channel sounding signal (S_(k)) 253, achannel estimate (H_(k)) 251, and a noise matrix (N_(k)) 257. Inparticular, the received signal X_(k)=S_(k)*H_(k)+N_(k), where S_(k),H_(k), and N_(k) are matrixes. In one embodiment, the channel estimateH_(k) 251 and the transmitted channel sounding signal S_(k) 253 may bein the form of: $\begin{matrix}{X_{k} = {{S_{k} \cdot H_{k}} + N_{k}}} \\{H_{k} = \begin{pmatrix}h_{k}^{({0,0})} & h_{k}^{({0,1})} & \cdots & h_{k}^{({0,{N - 1}})} \\h_{k}^{({1,0})} & h_{k}^{({1,1})} & \cdots & h_{k}^{({1,{N - 1}})} \\\vdots & \vdots & ⋰ & \vdots \\h_{k}^{({{M - 1},0})} & h_{k}^{({{M - 1},1})} & \cdots & h_{k}^{({{M - 1},{N - 1}})}\end{pmatrix}} \\{S_{k} = \begin{pmatrix}s_{k}^{({0,0})} & s_{k}^{({0,1})} & \cdots & s_{k}^{({0,{M - 1}})} \\s_{k}^{({1,0})} & s_{k}^{({1,1})} & \cdots & s_{k}^{({1,{M - 1}})} \\\vdots & \vdots & ⋰ & \vdots \\s_{k}^{({{L - 1},0})} & s_{k}^{({{L - 1},1})} & \cdots & s_{k}^{({{L - 1},{M - 1}})}\end{pmatrix}}\end{matrix}$From this signal-model, the zero-forcing (ZF) MIMO channel estimate isthen computed as:${\hat{H}}_{k} = {{\left( {S_{k}^{H} \cdot S_{k}} \right)^{- 1} \cdot S_{k}^{H} \cdot X_{k}} = {\frac{1}{M} \cdot S_{k}^{H} \cdot X_{k}}}$

If the long training symbol sequence is defined well, S_(k) ends upbeing a real scalar times a unitary matrix). In such case, the minimummean-square (MMSE) channel estimate is computed as: $\begin{matrix}{{\hat{H}}_{k} = {{\left( {{S_{k}^{H} \cdot S_{k}} + {\sigma_{\eta}^{2} \cdot I}} \right)^{- 1} \cdot S_{k}^{H} \cdot X_{k}} = {\rho \cdot S_{k}^{H} \cdot X_{k}}}} \\{\rho = \frac{1}{M + \sigma_{\eta}^{2}}}\end{matrix}$where, for simplicity, n_(k) is assumed to be individually identicallydistributed (i.i.d.) Gaussian and chosen to make a “good long trainingchoice.” Note that there is essentially no reason to perform MMSE vs.Zero Forcing (ZF) estimation for the sequences since we have S iscarefully chosen.

FIG. 13 is a diagram illustrating the transmission of a plurality ofpreambles 261-265 on a plurality of antennas (TX 1 through TX M) that iscompatible with the baseband processing modules of both FIGS. 7 and 8.In one embodiment, each preamble 261-265 includes a carrier detect (CD)field 267, 277, 287, a first channel sounding (CS M,1) 269, 279, and289, a signal field (SIG) 271, 281, 291, and L-1 remaining channelsoundings (CS M,L). In such an embodiment, the channel detect CD 267,277, 287, the first channel soundings 269, 279, 289, and the signalfield 271, 281, 291 may correspond to a short training sequence, a longtraining sequence, and a signal field of a legacy wireless protocol(e.g., IEEE 802.11a, b, and/or g).

According to this teaching of the present invention, preamble energy istransmitted from an IEEE 802.11n STA or AP on all tones, or nearly allof the tones, on all antennas, or nearly all of the antennas, for all Lsounding sequences. The energy sent from each of M antennas during eachof L soundings is 2s²/M when L=M. The total energy for matrix channelestimation is 2Ms² when L=M. Thus, the transmitted energy is M timesmore energy sent than when only a single tone is transmitted at anytime.

FIG. 14 is a diagram illustrating a transmission model of the frameformat of FIG. 12A. For this transmission format, in order to satisfybackwards compatibility issues and also to satisfy the requirements ofthe next generation channel estimation requirements, W is chosen suchthat W and W⁻¹ are simple to implement. Further, any beam forming issuesfrom MIMO transmitters (next generation devices) by [w₁₁ . . . w_(1M)]should be well-received by legacy 802.11a/g devices.

In this embodiment, a channel sounding (S_(k)) 253 is multiplied by aplurality of weighting factors (W_(k,m)) 68-72, wherein k corresponds tothe channel sounding number, which ranges from 1 to l, and m correspondsto the number of transmit antennas 82-86. The resulting weighted channelsoundings are converted to RF signals via the transmitters 68-72 andsubsequently transmitted via the antennas 82-86. In such an embodiment,a weighting factor matrix may be as follows: $\begin{bmatrix}w_{11} & w_{12} & \cdots & w_{1M} \\w_{21} & w_{22} & \cdots & w_{2M} \\\vdots & \vdots & ⋰ & \vdots \\w_{L1} & w_{L2} & \cdots & w_{L\quad M}\end{bmatrix}s_{k}$

With transmissions occurring on all antennas at all times, nulls may beformed. The nulls may be compensated for by selecting a weight sequencethat acts as a beam former such that the nulls are steered in particulardirections. For example, for the case of the vector w₁=[11] (one row ofthe previous slide's W matrix for a 2 TX case), nulls would be steeredin the directions −90° and +90°. Thus, certain directions aredisadvantaged vs. others at a single-input receiver of a legacy WLANdevice.

According to the present invention, a different complex weight isapplied to each subcarrier on M−1 of the transmit antennas. This forms adifferent beam pattern on each subcarrier and results in less power andcapacity loss in the worst directions.

FIG. 15 is a diagram illustrating the manner in which a preamble of theframe format of FIG. 12A is formed for a generalized next generationMIMO transmitter and particularly for a two antenna next generation MIMOtransmitter. In this illustration, two preambles are generated: one foreach active antenna. The first preamble 311, which is transmitted by thefirst antenna, includes a double guard interval (GI2) 313, a firstchannel sounding (CS 0,0) 315, a second channel sounding (CS 0,1) 317, aguard interval (GI) 319, a signal field (SIG) 321, another guardinterval (GI) 323, and a third channel sounding (CS 0,2) 325. The secondpreamble 327, which is transmitted by the second antenna, includes adouble guard interval (GI2) 329, a first channel sounding (CS 1,0) 331,a second channel sounding (CS 1,1) 333, a guard interval (GI) 335, asignal field (SIG) 337, another guard interval (GI) 339, and a thirdchannel sounding (CS 1,2) 341.

In this embodiment, the following may be used for the various channelsoundings:s₀₁=s₀₀s _(10,k) =−s _(00,k) ·e ^(i·0) ^(k)s₁₁=s₁₀s₀₂=s₀₀s _(12,k) =s _(00,k) ·e ^(i·0) ^(k)From these channel soundings, the weighting factor may be applied asfollows: $\begin{matrix}{S_{k} = {\begin{bmatrix}s_{10,k} & s_{11,k} \\s_{20,k} & s_{21,k}\end{bmatrix} = {s_{00,k} \cdot \begin{bmatrix}1 & {- 1} \\1 & 1\end{bmatrix} \cdot \begin{bmatrix}1 & 0 \\0 & {\mathbb{e}}^{{\mathbb{i}} \cdot \theta_{k}}\end{bmatrix}}}} \\{= \begin{bmatrix}s_{00,k} & {{- s_{00,k}} \cdot {\mathbb{e}}^{{\mathbb{i}} \cdot \theta_{k}}} \\s_{00,k} & {s_{00,k} \cdot {\mathbb{e}}^{{\mathbb{i}} \cdot \theta_{k}}}\end{bmatrix}}\end{matrix}$where the first digital of the subscript of a channel soundingcorresponds to the number of antennas, the second digit corresponds tothe number of symbols, and the k corresponds to the number of channelsoundings. For example, S_(10,k) corresponds to the first symboltransmitted on the first antenna for the kth channel sounding.

To obtain a different beam pattern for each subcarrier, the following isapplied:${\theta_{k} = {\pi \cdot {k/6}}},{k = {{- \frac{N_{subcarriers}}{2}}\quad\ldots\quad\frac{N_{subcarriers}}{2}}}$

FIG. 16 is a diagram illustrating the manner in which a preamble of theframe format of FIG. 12A is formed for a three antenna next generationMIMO transmitter. In this illustration, three preambles are generated:one for each active antenna. The first preamble 351, which istransmitted by the first antenna, includes a double guard interval (GI2)353, a first channel sounding (CS 0,0) 355, a second channel sounding(CS 0,1) 357, a guard interval (GI) 359, a signal field (SIG) 361,another guard interval (GI) 363, a third channel sounding (CS 0,2) 365,a third guard interval (GI) 367, and a fourth channel sounding (CS 0,3)369. The second preamble 371, which is transmitted by the secondantenna, includes a double guard interval (GI2) 373, a first channelsounding (CS 1,0) 375, a second channel sounding (CS 1,1) 377, a guardinterval (GI) 379, a signal field (SIG) 381, another guard interval (GI)383, a third channel sounding (CS 1,2) 385, a third guard interval (GI)387, and a fourth channel sounding (CS 1,3) 389. The third preamble 391,which is transmitted by the third antenna, includes a double guardinterval (GI2) 393, a first channel sounding (CS 2,0) 395, a secondchannel sounding (CS 2,1) 397, a guard interval (GI) 399, a signal field(SIG) 401, another guard interval (GI) 403, a third channel sounding (CS2,2) 405, a third guard interval (GI) 407, and a fourth channel sounding(CS 2,3) 409.

For the various channel soundings, the weighting factor matrix may beapplied as follows: $S_{k} = {\begin{bmatrix}s_{10,k} & s_{11,k} & s_{12,k} \\s_{20,k} & s_{21,k} & s_{22,k} \\s_{30,k} & s_{31,k} & s_{32,k}\end{bmatrix} = \begin{bmatrix}s_{00,k} & {s_{00,k} \cdot {\mathbb{e}}^{{\mathbb{i}} \cdot \theta_{k}}} & {s_{00,k} \cdot {\mathbb{e}}^{{\mathbb{i}} \cdot \phi_{k}}} \\s_{00,k} & {s_{00,k} \cdot {\mathbb{e}}^{{\mathbb{i}} \cdot {({\theta_{k} - \frac{4 \cdot \pi}{3}})}}} & {s_{00,k} \cdot {\mathbb{e}}^{{\mathbb{i}} \cdot {({\phi_{k} - \frac{2 \cdot \pi}{3}})}}} \\s_{00,k} & {s_{00,k} \cdot {\mathbb{e}}^{{\mathbb{i}} \cdot {({\theta_{k} - \frac{2 \cdot \pi}{3}})}}} & {s_{00,k} \cdot {\mathbb{e}}^{{\mathbb{i}} \cdot {({\phi_{k} - \frac{4 \cdot \pi}{3}})}}}\end{bmatrix}}$To obtain a different beam pattern for each subcarrier, the following isapplied:θ_(k) =π·k/6φ_(k)=π·(k+4)/6

FIG. 17 is a diagram illustrating the manner in which a preamble of theframe format of FIG. 12A is formed for a four antenna next generationMIMO transmitter. In this illustration, four preambles are generated:one for each active antenna. The first preamble 411, which istransmitted by the first antenna, includes a double guard interval (GI2)413, a first channel sounding (CS 0,0) 415, a second channel sounding(CS 0,1) 417, a guard interval (GI) 419, a signal field (SIG) 421,another guard interval (GI) 423, a third channel sounding (CS 0,2) 425,a third guard interval (GI) 427, a fourth channel sounding (CS 0,3) 429,a guard interval (GI) 431, and a fifth channel sounding (CS 0,4) 435.The second preamble 441, which is transmitted by the second antenna,includes a double guard interval (GI2) 443, a first channel sounding (CS1,0) 445, a second channel sounding (CS 1,1) 447, a guard interval (GI)449, a signal field (SIG) 451, another guard interval (GI) 453, a thirdchannel sounding (CS 1,2) 455, a third guard interval (GI) 457, a fourthchannel sounding (CS 1,3) 459, a guard interval (GI) 461, and a fifthchannel sounding (CS 1,4) 465. The third preamble 471, which istransmitted by the third antenna, includes a double guard interval (GI2)473, a first channel sounding (CS 2,0) 475, a second channel sounding(CS 2,1) 477, a guard interval (GI) 479, a signal field (SIG) 481,another guard interval (GI) 483, a third channel sounding (CS 2,2) 485,a third guard interval (GI) 487, a fourth channel sounding (CS 2,3) 489,a guard interval (GI) 491, and a fifth channel sounding (CS 2,4) 495.The fourth preamble 501, which is transmitted by the fourth antenna,includes a double guard interval (GI2) 503, a first channel sounding (CS3,0) 505, a second channel sounding (CS 3,1) 507, a guard interval (GI)509, a signal field (SIG) 511, another guard interval (GI) 513, a thirdchannel sounding (CS 3,2) 515, a third guard interval (GI) 517, a fourthchannel sounding (CS 3,3) 519, a guard interval (GI) 521, and a fifthchannel sounding (CS 3,4) 525.

For the embodiment of FIG. 17,θ_(k) =π·k/6φ_(k)=π·(k+2)/6ψ_(k)=π·(k+4)/6

With the operations of FIGS. 15-17, θ_(k), φ_(k), and ψ_(k) are set toform a different beam pattern on each subcarrier. Because more energy istransmitted, better channel estimates may be determined by nextgeneration 802.11n devices. Further, with this signal format, simpleZero Forcing (ZF) or MMSE channel estimation may be performed by thenext generation receivers.

Such channel estimation operations may be performed by applying thefollowing matrices for the two antenna, three antenna, and four antennacases, respectively. $\begin{matrix}{W_{T} = {\left. \begin{bmatrix}{+ 1} & {- 1} \\{+ 1} & {+ 1}\end{bmatrix}\Rightarrow W_{T}^{- 1} \right. = {\frac{1}{2}\begin{bmatrix}{+ 1} & {+ 1} \\{- 1} & {+ 1}\end{bmatrix}}}} \\{W_{T} = \left. \begin{pmatrix}1 & 1 & 1 \\1 & \frac{{- 1} - {{\mathbb{i}} \cdot \sqrt{3}}}{2} & \frac{{- 1} + {{\mathbb{i}} \cdot \sqrt{3}}}{2} \\1 & \frac{{- 1} + {{\mathbb{i}} \cdot \sqrt{3}}}{2} & \frac{{- 1} - {{\mathbb{i}} \cdot \sqrt{3}}}{2}\end{pmatrix}\Rightarrow \right.} \\{W_{T}^{- 1} = {\frac{1}{3} \cdot \begin{pmatrix}1 & 1 & 1 \\1 & \frac{{- 1} + {{\mathbb{i}} \cdot \sqrt{3}}}{2} & \frac{{- 1} - {{\mathbb{i}} \cdot \sqrt{3}}}{2} \\1 & \frac{{- 1} - {{\mathbb{i}} \cdot \sqrt{3}}}{2} & \frac{{- 1} + {{\mathbb{i}} \cdot \sqrt{3}}}{2}\end{pmatrix}}} \\{W_{T} = \left. \begin{bmatrix}{- 1} & {+ 1} & {+ 1} & {+ 1} \\{+ 1} & {- 1} & {+ 1} & {+ 1} \\{+ 1} & {+ 1} & {- 1} & {+ 1} \\{+ 1} & {+ 1} & {+ 1} & {- 1}\end{bmatrix}\Rightarrow \right.} \\{W_{T}^{- 1} = {\frac{1}{4}\begin{bmatrix}{- 1} & {+ 1} & {+ 1} & {+ 1} \\{+ 1} & {- 1} & {+ 1} & {+ 1} \\{+ 1} & {+ 1} & {- 1} & {+ 1} \\{+ 1} & {+ 1} & {+ 1} & {- 1}\end{bmatrix}}}\end{matrix}$

Using these techniques, in a first embodiment, the channel may beestimated with prior knowledge of the per-subcarrier beamformingcoefficients and then these coefficients do not need to be applied tothe remaining transmitted symbols. This embodiment provides theadvantage that no extra multiplications are required on the transmitterside, as the LTRN sequence may simply be looked up in a table.

With a second embodiment, the channel may be estimated without knowledgeof the per-subcarrier beamforming coefficients. With this embodiment,the coefficients must be applied to the remaining transmitted symbols.An advantage of this embodiment is that the receiver channel estimationis simplified (fewer multiplies), but the transmitter performsadditional multiplications.

With the first embodiment, the following equations apply (using earliernotation and L=M:${\hat{H}}_{k} = {\frac{1}{M} \cdot s_{00,k}^{*} \cdot W_{B,k}^{H} \cdot W_{T}^{H} \cdot X_{k}}$$W_{B,k} = {{diag}\left( \begin{bmatrix}1 & {\mathbb{e}}^{\frac{{\mathbb{i}} \cdot \pi \cdot l_{1}^{\prime}}{6}} & \ldots & {\mathbb{e}}^{\frac{{\mathbb{i}} \cdot \pi \cdot l_{M - 1}}{6}}\end{bmatrix} \right)}$

With the second embodiment, the following equation applies (usingearlier notation and L=M:${\hat{H}}_{k} = {\frac{1}{M} \cdot s_{00,k}^{*} \cdot W_{T}^{H} \cdot X_{k}}$

Further refinement of the channel estimate is possible by duplicatingthe entire length-M sequence p times. The refinement may be made bysimple averaging. The overhead is identical to the single activetransmitter method described on slide 10, but the performance is farsuperior.

For the backward-compatible preamble case, in which the number of longtraining symbols is M+1, the longer sequence would consist of p*M+1 longtraining symbols. There are p identical blocks of M symbols, and thefirst and second symbols on each antenna are identical.

FIG. 18 is a block diagram illustrating the manner in which legacy andnext generation WLAN devices interpret information contained in a headerof a frame that is backwards compatible. To reduce receivercomputational complexity, the transmit antenna and preamble 531configuration needs to be encoded in the legacy SIGNAL field 533. Thealternative is to have the receiver compute 4 different channelestimates and then select the antenna/preamble configuration 535 thatresults in a parity bit match and only legal values in the SIGNAL2 (MIMOextensions) field.

According to one aspect of the present invention, if the Reserved bit inthe SIGNAL field is set, the Rate bits 537 are re-interpreted using the“MIMO interpretation” 539. For MIMO receivers, with which the rate isdetermined to be fixed, the Rate bits no longer specify the actual rate.Instead, they specify a dummy rate which, in combination with the lengthfield uniquely identify the length of the frame in symbols and the TXantenna/preamble configuration.

For example, for 54 Mbps, there are 27 possible numbers of bytes in thelength field that yield the same frame duration in symbols. These 27possibilities can encode the TX antenna/preamble configuration. In thefollowing Table 1, three encodings are shown. Note that we may simply beable to use the “6 Mbps” Rate to uniquely specify all lengths and txantenna/preamble configurations. In this case, the other Rate codeswould not be used if the Reserved bit is set. The only disadvantage isthat we lose the ability to encode other preamble choices. TABLE 1 RateField Interpretation 802.11a interpretation MIMO interpretation Ratebits (reserved bit = 0) (reserved bit = 1) 1101  6 Mbps length mod 3 = 1=> 2 TX antennas length mod 3 = 2 => 3 TX antennas length mod 3 = 3 => 4TX antennas 1111  9 Mbps N/A 0101 12 Mbps length mod 6 = 1 => 2 TXantennas length mod 6 = 2 => 3 TX antennas length mod 6 = 3 => 4 TXantennas 0111 18 Mbps length mod 9 = 1 => 2 TX antennas length mod 9 = 2=> 3 TX antennas length mod 9 = 3 => 4 TX antennas 1001 24 Mbps lengthmod 12 = 1 => 2 TX antennas length mod 12 = 2 => 3 TX antennas lengthmod 12 = 3 => 4 TX antennas 1011 36 Mbps length mod 18 = 1 => 2 TXantennas length mod 18 = 2 => 3 TX antennas length mod 18 = 3 => 4 TXantennas 0001 48 Mbps length mod 24 = 1 => 2 TX antennas length mod 24 =2 => 3 TX antennas length mod 24 = 3 => 4 TX antennas 0011 54 Mbpslength mod 27 = 1 => 2 TX antennas length mod 27 = 2 => 3 TX antennaslength mod 27 = 3 => 4 TX antennas

FIG. 19 is a graph illustrating the frequency characteristics of variousspectral masks of current IEEE 802.11a, b, and g masks 541 and aplurality of next generation MIMO signal masks according to the presentinvention 543, 545, 547. The requirements of the next generationinterface may include an effective data rate. One such effective datarate requires 100 Mbps throughput in 20 MHz Channel. Of course, lowerdata rates provide a longer reach between devices supporting the signalformat. For example, the graph includes a horizontal axis in MHzrepresenting an offset from the center frequency of a 20 MHz channel anda vertical axis in decibels (dB). In this example, spectral mask 543 maybe for each path of a 20 MHz channel with 48 subcarriers of 64subcarriers being used to carry data. Spectral mask 545 may be for eachpath of a 20 MHz channel with 52 subcarriers of 64 subcarriers beingused to carry data. Spectral mask 547 may be for each path of a 20 MHzchannel with 54 subcarriers of 64 subcarriers being used to carry data.

For 100 Mbps throughput, 130 Mbps is typically required at the PHY ofthe servicing devices. Other requirements may include for example, aparticular frame format, e.g., 4096 byte frames, a number of frames in aburst, e.g., 10 frames in a burst (802.11e TXOP bursting), and supportof features to prevent inter-device conflicts, e.g., RTS/CTS. To meetthese various requirements, MIMO devices are contemplated.

For a two TX path MIMO device/system, additional characteristics arealso required to be selected. For example, a system may be specifiedhaving the following various characteristics:

-   -   2 TX Path MIMO    -   48 subcarriers*2 paths/4.0 microseconds=24 Msym/sec    -   130/24=5.416 bits/symbol    -   64 QAM—rate 0.9 code    -   128 QAM—rate 0.7737 code    -   256 QAM—rate 0.677 code    -   Preferable to add some more carriers, if possible    -   Alternating pilot subcarriers on different TX streams

Each of these combinations of characteristics meets the requirements forthe next generation system. These combinations may be furthercharacterized as follows:

Option #1—Larger Constellation

50 subcarriers active per stream (2 pilots per stream alternating)

(−21,−7),(−21,7),(−21,21),(−7,7),(−7,21),(7,21) on path 1

(7,21),(−7,21),(−7,7),(−21,21),(−21,7),(−21,−7) on path 2

128 QAM

Rate 0.742→0.75 code rate

Option #2—Higher Rate Code

52 subcarriers active per stream (2 pilots per stream alternating+2additional tones at bins±27)

64 QAM

Rate 5/6 code

Option #3—Shorter Cyclic Prefix

52 data subcarriers per stream

cut cyclic prefix in half (0.4 microsecond cyclic prefix→3.6 microsecondsymbol)

64 QAM

rate 3/4 code

Each of these combinations provides various advantages and disadvantageswhile meeting the requirements set forth, each in the context of a twoantenna MIMO system. With the MIMO signal format of the next generationWLAN system, smaller constellations and lower code rates may be employedwhile still providing a longer reach and an equal or greater data rate,as compared to legacy IEEE 802.11a/g systems. For example, a 2by 2 MIMOsystem using a 20 MHz channel supports 12, 24, and 48 Mbps rates.Further, a 2 by 2 MIMO system using a 40 MHz channel supports 27 and 54Mbps. Each of these next generation systems provides operationaladvantages over legacy IEEE 802.11a/g systems. Further, with a 4 by 4MIMO system, higher data rates are supported using comparable codingrates to legacy IEEE 802.11a/g systems. In particular, a 4 by 4 MIMOsystem using a 3/4 coding rate (as in IEEE 802.11a/g) in a 20 MHzchannel supports 192 Mbps. Further, a 4 by 4 MIMO system using a 3/4coding rate (as in IEEE 802.11a/g) in a 40 MHz channel supports 486Mbps.

As one of average skill in the art will appreciate, the term“substantially” or “approximately”, as may be used herein, provides anindustry-accepted tolerance to its corresponding term. Such anindustry-accepted tolerance ranges from less than one percent to twentypercent and corresponds to, but is not limited to, component values,integrated circuit process variations, temperature variations, rise andfall times, and/or thermal noise. As one of average skill in the artwill further appreciate, the term “operably coupled”, as may be usedherein, includes direct coupling and indirect coupling via anothercomponent, element, circuit, or module where, for indirect coupling, theintervening component, element, circuit, or module does not modify theinformation of a signal but may adjust its current level, voltage level,and/or power level. As one of average skill in the art will alsoappreciate, inferred coupling (i.e., where one element is coupled toanother element by inference) includes direct and indirect couplingbetween two elements in the same manner as “operably coupled”. As one ofaverage skill in the art will further appreciate, the term “comparesfavorably”, as may be used herein, indicates that a comparison betweentwo or more elements, items, signals, etc., provides a desiredrelationship. For example, when the desired relationship is that signal1 has a greater magnitude than signal 2, a favorable comparison may beachieved when the magnitude of signal 1 is greater than that of signal 2or when the magnitude of signal 2 is less than that of signal 1.

The preceding discussion has presented a method and apparatus forupdating a channel estimation based on payload of a frame. As one ofaverage skill in the art will appreciate, other embodiments may bederived from the present discussion without deviating from the scope ofthe claims.

1. A method for transmitting high rate data within a multiple inputmultiple output (MIMO) wireless local area network (WLAN), the methodcomprises: determining a data transmission rate; when the datatransmission rate is between a first data rate and a second data rate,enabling two transmission paths; for each of the two transmission paths,determining at least one of: level of constellation; number of datasubcarriers; rate code; and cyclic prefix duration.
 2. The method ofclaim 1, wherein the enabling the two transmission paths comprises:activating two antennas for a 20 MHz channel.
 3. The method of claim 1,wherein the level of constellation comprises at least one of: 64Quadrature Amplitude Modulation (QAM); 128 QAM; and 256 QAM.
 4. Themethod of claim 3, wherein the determining the number of datasubcarriers when the level of constellation is 64 QAM comprises at leastone of: selecting 52 subcarriers of 64 subcarriers for carrying data;and selecting 54 subcarriers of 64 subcarriers for carrying data.
 5. Themethod of claim 4 comprises, when 52 subcarriers are selected,:utilizing alternating two pilot subcarriers per each of the twotransmission paths; and selecting a rate 5/6 code.
 6. The method ofclaim 4 comprises, when 54 subcarriers are selected,: utilizingalternating two pilot subcarriers per each of the two transmissionpaths; and utilizing subcarriers +27 and −27 for carrying data.
 7. Themethod of claim 4 comprises, when 52 subcarriers are selected,: reducingcyclic prefix; and selecting a rate 3/4 code.
 8. The method of claim 4further comprises: adjusting a spectral mask based on the number ofsubcarriers selected.
 9. The method of claim 1 further comprises: whenthe data transmission rate is less than the first data rate due todistance, enabling the two transmission paths to utilize a 20 MHzchannel with code rates less than ¾ for 12, 24, or 48 Mbps (Mega bitsper second) data rates or enabling the two transmission paths to utilizea 40 MHz with code rates less than ¾ for 27 or 54 Mbps data rates. 10.The method of claim 1 further comprises: when the data transmission rateis greater than the second data rate, enabling four transmission pathsutilizing a 20 MHz channel and a code rate of ¾ for 192 Mbps data rateor enabling the four transmission paths utilizing a 40 MHz channel and acode rate of ¾ for 486 Mbps data rate.
 11. A method for supporting highdata rate WLAN communications comprising: determining a bandwidth ofoperation; determining a required data throughput rate; selecting anumber of antennas for use in a Multiple Input Multiple Output (MIMO)baseband signal format; selecting a constellation; and operating a MIMOWLAN transceiver according to the bandwidth of operation, the number ofantennas, and the constellation to meet the required data throughputrate.
 12. The method of claim 11: further comprising determining acoding rate of operation; and wherein the MIMO WLAN transceiver furtheroperates according to the coding rate of operation.
 13. The method ofclaim 11, wherein the MIMO WLAN transceiver supports an OrthogonalFrequency Division Multiplexed (OFDM) baseband signal format.
 14. Themethod of claim 13, wherein the OFDM baseband signal format includes 50subcarriers for carrying data within the bandwidth of operation perantenna.
 15. The method of claim 13, wherein the OFDM baseband signalformat includes 52 subcarriers within the bandwidth of operation perantenna.
 16. The method of claim 11: further comprising determining areduced cyclic prefix format; and wherein the MIMO WLAN transceiverfurther operates according to the reduced cyclic prefix format.
 17. Atransmitter comprises: baseband processing module; and a plurality ofradio frequency (RF) transmitters, wherein the processing modulefunctions to: determine a data transmission rate; when the datatransmission rate is between a first data rate and a second data rate,enable two of the plurality of RF transmitters; for each of the two ofthe plurality of RF transmitters, determining at least one of: level ofconstellation; number of data subcarriers; rate code; and cyclic prefixduration.
 18. The transmitter of claim 17, wherein the basebandprocessing module functions to enable the two of the plurality of RFtransmitters by: activating the two of the plurality of RF transmittersfor a 20 MHz channel.
 19. The transmitter of claim 17, wherein the levelof constellation comprises at least one of: 64 Quadrature AmplitudeModulation (QAM); 128 QAM; and 256 QAM.
 20. The transmitter of claim 19,wherein the determining the number of data subcarriers when the level ofconstellation is 64 QAM comprises at least one of: selecting 52subcarriers of 64 subcarriers for carrying data; and selecting 54subcarriers of 64 subcarriers for carrying data.
 21. The transmitter ofclaim 20, wherein the baseband processing module further functions to,when 52 subcarriers are selected,: utilize alternating two pilotsubcarriers per frame for each of the two of the plurality of RFtransmitters; and select a rate 5/6 code.
 22. The transmitter of claim20, wherein the baseband processing module further functions to, when 54subcarriers are selected,: utilize alternating two pilot subcarriers perframe for each of the two of the plurality of RF transmitters; andutilize subcarriers +27 and −27 per frame for carrying data.
 23. Thetransmitter of claim 20, wherein the baseband processing module furtherfunctions to, when 52 subcarriers are selected,: reduce cyclic prefix ofa frame; and select a rate 3/4 code.
 24. The transmitter of claim 20,wherein the baseband processing module further functions to: adjust aspectral mask based on the number of subcarriers selected.
 25. Thetransmitter of claim 17, wherein the baseband processing module furtherfunctions to: when the data transmission rate is less than the firstdata rate due to distance, enable the two of the plurality of RFtransmitters to utilize a 20 MHz channel with code rates less than ¾ for12, 24, or 48 Mbps (Mega bits per second) data rates or enabling the twoof the plurality of RF transmitters to utilize a 40 MHz with code ratesless than ¾ for 27 or 54 Mbps data rates.
 26. The transmitter of claim17, wherein the baseband processing module further functions to: whenthe data transmission rate is greater than the second data rate, enablefour of the plurality of RF transmitters utilizing a 20 MHz channel anda code rate of ¾ for 192 Mbps data rate or enabling the four of theplurality of RF transmitters utilizing a 40 MHz channel and a code rateof ¾ for 486 Mbps data rate.
 27. A transmitter comprises: basebandprocessing module; and a plurality of radio frequency (RF) transmitters,wherein the processing module functions to: determine a bandwidth ofoperation; determine a required data throughput rate; select a number ofthe plurality of RF transmitters for use in a Multiple Input MultipleOutput (MIMO) baseband signal format to produce selected RFtransmitters; select a constellation; and operate the selected RFtransmitters according to the bandwidth of operation, and theconstellation to meet the required data throughput rate.
 28. Thetransmitter of claim 27, wherein the baseband processing module furtherfunctions to: determine a coding rate of operation; and wherein theselect RF transmitters further operates according to the coding rate ofoperation.
 29. The transmitter of claim 27, wherein each of the selectedRF transmitters supports an Orthogonal Frequency Division Multiplexed(OFDM) baseband signal format.
 30. The transmitter of claim 29, whereinthe OFDM baseband signal format includes 50 subcarriers for carryingdata within the bandwidth of operation per each of the selected RFtransmitters.
 31. The transmitter of claim 29, wherein the OFDM basebandsignal format includes 52 subcarriers within the bandwidth of operationper each of the selected RF transmitters.
 32. The transmitter of claim27, wherein the baseband processing module further functions to:determine a reduced cyclic prefix format; and wherein the selected RFtransmitters further operate according to the reduced cyclic prefixformat.